This invention relates to receivers for many-carrier signals, for example OFDM (orthogonal frequency-division multiplex) signals, such as used in digital audio broadcasting (DAB) or digital sound broadcasting, and also in digital television.
The digital audio broadcasting system used in the United Kingdom, now known as mode 1 DAB, requires seven DAB ensembles (channels) each occupying 1.536 MHz within the overall frequency range 217.5 MHz to 230 MHz, a total bandwidth of only 12.5 MHz. The spacing between ensemble centres is 1712 KHz, of which 1536 kHz is taken up by the signal, so that the spacing between the top of one ensemble and the bottom of the next is only 176 kHz. The signal bandwidth of 1536 kHz arises from the use of 1536 separate carriers spaced at spacings of 1 kHz. The number 1536 is chosen as three-quarters of 2048, which equals 2 to the power of eleven, that being the minimum power of 2 which is greater than 1536.
A DAB receiver must be able to receive the desired DAB ensemble to which it is tuned in the presence of a number of interfering DAB signals occupying the adjacent spectrum. Of the adjacent signals, it is the nearest neighbour which is most difficult to reject because of the high ratio of signal bandwidth to edge spacing; the edge spacing is less than one-eighth of the signal bandwidth.
A typical DAB receiver of this type is shown in schematic block diagram form in FIG. 1 of the drawings. The receiver 100 has an antenna 112 feeding a radio frequency (RF) stage shown as an RF amplifier 114. The output of the RF amplifier is applied to a mixer 116 which receives a first local oscillator signal LO1 at a terminal 118. The mixer reduces the frequency of the received signal, typically at around 225 MHz, to an intermediate frequency of typically about 36 MHz. The output of the mixer 116 is applied to an IF bandpass filter 120 which passes the desired intermediate frequency in the region of 36 MHz. The output of the IF filter 120 is then applied to an I/Q demodulator circuit 122, which receives a second local oscillator signal LO2 at a terminal 124. The I/Q demodulator circuit reduces the signals to baseband frequency and also separates the in-phase (I) and quadrature phase (Q) components of the signal. The output 126 of the I/Q demodulator thus in fact comprises two signals, as indicated on FIG. 1, and the subsequent circuitry is duplicated for the two signals, as is well known.
The output 126 of the I/Q demodulator 122 is applied to an anti-alias bandpass filter 128, the characteristics of which are described in more detail below, and from the anti-alias filter 128 are applied to a sampler or analog-to-digital converter 130. The sampler 130 operates at 2.048 Ms/s (mega-samples per second), which is of course the same as the sample rate of the digital signals which were used to form the transmitted signal at the transmitter. In the sampler 130 the signals are now converted from analog form to digital form, and are then applied to a fast Fourier transform (FFT) circuit 132. The FFT generates a signal in the form of a sequence or series of symbol periods. The FFT has 2048 points which corresponds to the theoretical number of carriers with a sampler operating at 2.048 Ms/s, a carrier spacing of 1 kHz, and an active symbol period of 1 ms. In fact as noted above, only 1536 carriers are used, the remainder having a theoretical amplitude of zero.
To achieve this places considerable demands on the filter 128. This filter should have a pass-band extending to xc2x1768 kHz (half of 1536 kHz) but a cut-off frequency of xc2x11024 kHz (half of 2048 kHz). This is a sharp cut-off and is difficult to achieve.
The output 134 of the FFT is a time-based signal which is then processed using conventional receiver circuitry (not shown).
The circuit of FIG. 1 will be known to those skilled in the art, and further description thereof is not necessary.
Likewise, a corresponding transmitter will be known to those skilled in the art, and includes a 2048-point inverse FFT operating in the digital domain corresponding to the FFT 132 at the receiver. The inverse FFT receives a conventional time-based signal and converts it into a many-carrier signal for transmission.
FIG. 2 is a spectrum diagram showing three adjacent ensembles in the frequency spectrum. The numerical values are those appropriate to the DAB system described above, and are referred to the centre frequency of the central ensemble E which is taken to be zero. One ensemble E+1 is shown above this with positive values and another ensemble Exe2x88x921 is shown below it with negative values. The values are in kilohertz, but as the individual carriers are spaced by 1 kHz, they can equally be treated as a count of carriers. The amplitudes of the signals shown are purely arbitrary; they are shown for convenience of illustration with a slight peak at the centre of each ensemble but in theory the amplitudes should be flat.
It will be seen that each ensemble extends over 1536 carriers, and that the spacing between corresponding points on the ensembles is 1712 carriers.
FIG. 2 also shows, for the central ensemble, the positions where the sampling frequency and the inverse appear. These fall at xc2x12048 carriers. The values of half the sampling frequency, fs/2, which fall at xc2x11024 carriers, are also shown. The value of half the sampling frequency is, as is well known, the Nyquist limit. Frequencies which appear above half the sampling frequency can not be correctly represented by the sampling process.
These frequencies above half the sampling frequency, when subjected to sampling, give rise to spurious components in the sampled signal known as aliased components. The aliased components are commonly thought of as through the signals in the range fs/2 to fs were xe2x80x9creflectedxe2x80x9d about the frequency fs/2. Thus, a frequency which is a g Hz below the sampling frequency fs, that is a signal of frequency (fsxe2x88x92g) Hz, gives rise to an alias component of frequency of g Hz. This is correct for basebands signals, but for signals above baseband, correctly what happens is that the signals above fs/2 are translated downwards by a frequency shift equal to the sampling frequency fs. Such shift occurs in fact for all integral multiples of the sampling frequency, but only the first and most powerful need be considered in practice.
This is illustrated in FIG. 3, which shows just the central ensemble E of FIG. 2 and the ensemble E+1 above it. It also shows the aliased components which arise by down-shifting the upper ensemble E+1 by the sampling frequency. Those frequencies which arise in the range from above the Nyquist frequency, or half the sampling frequency, namely 1024 kHz, up to the top of the upper ensemble E+1, namely 2480 kHz, are moved downwards by 2048 kHz. The aliased components Exe2x88x921 now span the frequency range +1024 to +432. Of these the frequencies in the range of xe2x88x92768 to +432 fall within the band of the wanted ensemble E. These can not be rejected by simple frequency-selective filtering. The shifted frequencies are marked in the figure by cross-hatching.
The FFT circuit 132 expects only 1536 carriers out of a possible 2048, and thus inherently rejects energy in the frequency range 768 kHz to 1280 kHz. This upper limit equals the sampling frequency 2048 kHz minus the expected upper carrier frequency limit of 768 kHz. Within the wanted band, this includes the rejection regions marked R on FIG. 3.
We have thus appreciated that to cut out the interference components requires strong IF and anti-alias filtering in the filters 120 and 128 of FIG. 1, in order to reject the adjacent channel energy from ensemble E+1 before it reaches the analog-to-digital converter or sampler 130. To produce a sufficiently sharp cut-off may require for example a surface acoustic wave (SAW) filter for use as the filter 128. Such filters are expensive and lossy and may result in the partial loss of a number of carriers located towards the edges of the ensemble. Although the DAB transmission is very robust, nevertheless degrading the signal in this way may reduce the system margins available to combat other sources of degradation, e.g. multipath distortion or channel noise. Other types of filter may introduce less loss of signal but can introduce considerable group delay ripple into the signal. In summary, the filter requirements are quite difficult to meet without simultaneously adversely affecting the wanted signal. This problem arises independently with the IF filter 120 and with the anti-aliasing filter 128, though primarily with the latter.
This problem arises in the analog processing due to the difficulty of making adequately effective filters. It might therefore be thought that the problem could be solved by constructing the filter 128 in the digital domain rather than in the analog part of the circuit. That is to say, the filter 128 (or at least the greater part of its functionality) would be placed after the sampler 130, instead of ahead of it in the signal processing chain. This would of course require the use of a higher sampling rate than 2.048 Ms/s and a rate such as 4.096 Ms/s could conveniently be chosen. Subsequent to the filter, a downsampler would be included in order to reduce the sample rate to the value of 2.048 Ms/s which can be accepted by the FFT device 132.
While such an arrangement, using a digital filter, should produce theoretically improved rejection of the interfering signal, we have appreciated that it necessarily increases the signal processing requirements by a considerable amount.
It has been proposed by Muschallik, C., in xe2x80x9cImproving an OFDM Reception using an Adaptive Nyquist Windowingxe2x80x9d, IEEE Transactions on Consumer Electronics, Vol. 42, No. 3, August 1996, pages 259-269, (see also International Patent Application WO96/41458) to improve the sensitivity to frequency errors in an OFDM receiver by using a guard interval which is equal in length to the xe2x80x9cusefulxe2x80x9d part of the symbol period, instead of being only one-quarter as long or less, as is generally considered. The receiver FFT is then of double the usual length and uses the samples in the guard interval as well as those in the useful part of the symbol period. This increases the carrier to noise level by 3 dB. The sample rate is however unchanged.
United Kingdom Patent Specification GB-A-2,304,504 and U.S. Pat. No. 5,357,502 similarly propose to increase the length of the FFT, so as to accommodate more of the received signal.
The invention in its various aspects is defined in the independent claims below to which reference should now be made. Advantageous features of the invention are set forth in the appendant claims.
A first embodiment of the invention is described in more detail below, and takes the form of a DAB receiver designed for receiving a many-carrier signal with 1536 activate carriers as generated by an inverse fast Fourier transform (FFT ) with 2048 points. The receiver has the usual RF stage, an IF filter and a demodulator, followed for each of the complex baseband I and Q signals by an anti-aliasing filter, a sampler, and an FFT device. The FFT is a 4096-point FFT, and is thus twice as long as required, and gathers twice as many points during each active symbol period. That is, the sampling rate is doubled. However, we have found that this enables the construction of the IF filter and, more particularly, the anti-aliasing filter, to be substantially simplified.
A second embodiment is designed for use in a receiver where the processing takes place on real signals rather than quadrature demodulated signals. In this case the FFT is a 8192-point FFT, and the signal is reduced to a low IF frequency. xe2x80x9cLowxe2x80x9d here means less than a value of the order of the channel bandwidth, or at least less than a single-digit multiple of, up to ten times, the channel bandwidth.